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T H AT Corporation
FEATURES
* Pre-trimmed BlackmerTM VCA & RMS-level detector * Wide supply voltage range: 4.5V ~16V * Low supply current: 3.5 mA typ. (15V) * Wide dynamic range: 117 dB (VCA) 60 dB (RMS-level detector)
Pre-trimmed Analog Engine(R) IC
THAT 4305 APPLICATIONS
* Compressors & Limiters * Gates & Expanders * AGCs * Line-operated dynamics processors * De-Essers * Duckers * Mixers * Level indicators * Companding noise reduction systems
Description
The THAT4305 is a single-chip Analog Engine optimized for low-cost applications. It incorporates a high-performance Blackmer voltagecontrolled amplifier (VCA) and log-responding RMS-level sensor. The VCA and RMS detector are pre-trimmed at wafer stage to deliver low distortion without further adjustment. Available only in a small (QSOP) surfacemount package, the 4305 is aimed at line-operated audio applications such as compressor/limiters, gates, and other dynamic processors. The part normally operates from a split supply voltage up to 16Vdc, drawing only 3.5mA at 15V. This IC also works at supply voltages as low as 4.5V, making it useful in some battery-operated products as well. The 4305 was developed specifically for use in low-cost dynamics processors, drawing from THAT's long history and experience with such designs. Both VCA control ports and the detector input and output are available for the designer to connect as s/he sees fit. As a result, the part is extremely flexible and can be configured for a wide range of applications including singleand multi-band companders, digital overload protectors, voltage-controlled faders, level indicators, etc. What really sets the 4305 apart from other manufacturers' offerings is the transparent sound of its Blackmer VCA, coupled with its accurate true-RMS level detector. This makes the IC useful in a wide range of analog audio products.
NC
VCA IN
NC
VCA OUT
EC-
EC+
NC
VCC
16
15
14 VCA
13
12
11
10
9
IN
OUT
EC+ EC-
IN
RMS
CT
OUT
1
NC
2
RMS IN
3
NC
4
CT
5
RMS OUT
6
GND
7
NC
8
VEE
Pin Name No Connection RMS IN No Connection CTIME RMS OUT GND NC Vee VCC No Connection EC+ ECVCA OUT No Connection VCA IN No Connection
Pin Number 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16
Table 1. THAT 4305 pin assignments Package 16 pin QSOP Order Number 4305Q16-U
Figure 1. THAT4305 equivalent block diagram
Table 2. Ordering Information
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THAT4305 Pre-trimmed Analog Engine
SPECIFICATIONS Absolute Maximum Ratings 1
Operating Temperature Range (TOP) Junction Temperature (TJ) Power Dissipation (PD) at TA=85 C -40 to +85 C -40 to +125 C 400mW Supply Voltages (VCC, VEE) VCA Control Voltage Storage Temperature Range (TST) 18V 0.6 V -40 to +125 C
Electrical Characteristics 2
Parameter Symbol Conditions Min Typ Max Units
Power Supply
Positive Supply Voltage Negative Supply Voltage Supply Current ICC IEE ICC IEE VCC VEE Referenced to GND Referenced to GND No Signal VCC=+15V, VEE= -15V VCC=+15V, VEE= -15V VCC=+5V, VEE= -5V VCC=+5V, VEE= -5V +4.5 -4.5 +16 -16 V V
3.5 -3.5 2 -2
5 -5
mA mA mA mA
Voltage Controlled Amplifier (VCA)
Max. I/O Signal Current VCA Gain Range Gain at 0V Control Gain-Control Constant Gain-Control Tempco G0 EC+/Gain (dB) EC/TCHIP EC+ = EC- = 0V -60 dB < gain < +60 dB Ref TCHIP=27C ROUT = 20 k 0 dB gain +15 dB gain +30 dB gain 0 dB gain 22Hz~22kHz, RIN=ROUT=20 k Total Harmonic Distortion THD VIN= -5dBV, 1kHz, EC+ = EC- = 0V -97.5 0.07 -95 0.15 dBV % iIN(VCA) + iOUT(VCA) -60 -1.0 0 6.2 +0.33 1.8 +60 +1.0 mApeak dB dB mV/dB %/C
Output Offset Voltage Change3 VOFF(OUT)
-
1 3 10
15 30 50
mV mV mV
Output Noise
eN(OUT)
RMS level detector
Output Voltage at Reference iIN eO(0) Output Error at Input Extremes eO(RMS)error iIN = 7.5 A RMS iIN = 200 nA RMS iIN = 200 A RMS Scale Factor Match to VCA -20 dB < VCA gain < +20 dB 1 a< iIN(RMS) < 100 A .95 1 1.05 -9 0 1 1 +9 3 3 mV dB dB
1. If the devices are subjected to stress above the Absolute Maximum Ratings, permanent damage may result. Sustained operation at or near the Absolute Maximum Ratings conditions is not recommended. In particular, like all semiconductor devices, device reliability declines as operating temperature increases. 2. Unless otherwise noted, TA=25C, VCC=+15V, VEE= -15V. 3. Reference is to output offset with -60dB VCA gain.
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Electrical Characteristics (con't) 2
Parameter Rectifier Balance Timing Current Filtering Time Constant Output Tempco Load Resistance Capacitive Load IT EO/TCHIP RL CL Ref TCHIP = 27 C -250mV < VOUTRMS< +250mV 2 150 Symbol Conditions 7.5A DCIN Min Typ 1 7.5 3467 X CTIME +0.33 Max 3 Units dB A s %/C k pf
Package Characteristics
Parameter Symbol Conditions Min Typ Max Units
Surface Mount Package
Type Thermal Resistance Thermal Resistance See Fig. 23 for dimensions 16 Pin QSOP 105 40 C/W C/W
JC JA
SO package in ambient SO package soldered to board
Environmental Regulation Compliance Soldering Reflow Profile
Complies with RoHS requirements JEDEC JESD22-A113-D (250 C)
C5 100p VCA In C2 10u
+15V R4 6k8 R3 20k
15 11 Ec+ 9 Vcc 13 VCA In VCA Out Gnd Vee 6 Ec8 12
R2 20k RMS In C1 C3 10u 22p NPO VCA Out U2 R1 5k1
2
U1B THAT4305
RMS In RMS Out 5 CT 4
RMS Out R5 2k
+15V C4 10u
C TIME 10u
Control Voltage
-15V U1A THAT4305
Figure 2. Simplified application circuit
Theory of Operation
The THAT 4305 Dynamics Processor combines THAT Corporation's proven exponentially controlled BlackmerTM Voltage-Controlled Amplifier (VCA) and log-responding RMS-Level Detector building blocks in a small package optimized for low cost designs. The part is fabricated using a proprietary, fully complementary, dielectric-isolation process. This process produces very high-quality bipolar transistors (both NPNs and PNPs) with unusually low collectorsubstrate capacitances. The 4305 takes advantage of these devices to deliver wide bandwidth and excellent audio performance while consuming very low
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THAT4305 Pre-trimmed Analog Engine
current and operating over a wide range of power supply voltages. For details of the theory of operation of the VCA and RMS Detector, we refer the interested reader to THAT Corporation's data sheets on the 2180-Series VCAs and the 2252 RMS Level Detector. Theory of the interconnection of exponentially controlled VCAs and log-responding level detectors is covered in THAT Corporation's application note AN101A, The Mathematics of Log-Based Dynamic Processors.
The VCA's noise performance varies with gain in a predictable way, but due to the way internal bias currents vary with gain, noise at the output is not strictly the product of a static input noise times the voltage gain commanded. At large attenuation, the noise floor is usually limited by the input noise of the output op-amp and its feedback resistor. At 0 dB gain, the noise floor of ~ -97.5 dBV is the result of the VCA's output noise current, converted to a voltage by the typical 20k I-V converter resistor (R2 in Figure 2). In the vicinity of 0 dB gain, the noise increases more slowly than the gain: approximately 5 dB noise increase for every 10 dB gain increase. Finally, as gain approaches 30 dB, output noise begins to increase directly with gain. While the 4305's VCA circuitry is very similar to that of the THAT 2180 Series VCAs, there are several important differences, as follows. 1. Supply current for the 4305 VCA depends on the supply voltage. At 5 V, approximately 800 uA is available for the sum of input and output signal currents. This increases to about 1.8 mA at 15 V. (Compare this to ~1.8 mA for a 2180 Series VCA when biased as recommended.) 2. The SYM control port (similar to that on the 2180 VCA) is not brought out to an external pin; it is driven from an internally trimmed current generator. 3. The control-voltage constant is approximately 6.2 mV/dB, due primarily to the higher internal operating temperature of the 4305 compared to that of the 2180 Series.
The VCA - in Brief
The VCA in the 4305 is based on THAT Corporation's highly successful complementary log-antilog gain cell topology (the BlackmerTM VCA) as used in THAT 2180-Series IC VCAs. VCA symmetry is trimmed during wafer probe for minimum distortion. No external adjustment is allowed. Input signals are currents in the VCA's VCAIN pin (pin 15). This pin is a virtual ground with a small dc offset, so in normal operation an input voltage is converted to input current via an appropriately sized resistor (R3 in Figure 2). Because the dc current associated with dc offsets present at the input pin plus any dc offset in the preceding stages will be modulated by gain changes (thereby becoming audible as thumps), the input pin is normally ac-coupled. This blocks such offset currents and reduces dc offset variation with gain. The VCA output signal, VCAOUT (pin 13), is also a current, inverted with respect to the input current. In normal operation, the output current is converted to a voltage via an external op-amp, where the current-to-voltage conversion ratio is determined by the feedback resistor connected between the op-amp's output and its inverting input (R2 in Figure 2). The resulting signal path through the VCA plus op-amp is noninverting. The VCA gain is controlled by the voltage applied between EC+ (pin 11) and EC- (pin 12). Note that any unused control port should be connected to ground (as EC+ is in Figure 2). The gain (in decibels) is proportional to (EC+ - EC-). The constant of proportionality is 6.2 mV/dB for the voltage at EC+ (relative to EC-). Note that neither EC+ or EC- should be driven more than 0.6 V away from ground.
The RMS Detector - in Brief
The 4305's detector computes RMS level by rectifying input current signals, converting the rectified current to a logarithmic voltage, and applying that voltage to a log-domain filter. The output signal is a dc voltage proportional to the decibel-level of the RMS value of the input signal current. Some ac component (at twice the input frequency plus higherorder even harmonics) remains superimposed on the dc output. The ac signal is attenuated by a log domain filter, which constitutes a single-pole rolloff with cutoff determined by an external capacitor (C4 in Figure 2).
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The rectifier is balanced to within 3 dB, so a small amount of fundamental (and higher odd-order harmonics) ripple can be present at the detector output. By design, this ripple contributes less total ripple than the even-order products that are naturally and inevitably present at the output of a perfectly balanced detector. As in the VCA, input signals are currents to the RMSIN pin (pin 2). This input is a virtual ground, so a resistor (R1 in Figure 2) is normally used to convert input voltages to the desired current. The level detector is capable of accurately resolving signals well below 10 mV (with a 5 k input resistor). However, if the detector is to accurately track such low-level signals, ac coupling (C1 in Figure 2) is required to prevent dc offsets from causing a dc current to flow in the detector's input, which would obscure low-level ac signal currents. The log-domain filter cutoff frequency is usually placed well below the frequency range of interest. For an audio-band detector, a typical value would be 5 Hz, or a 32 ms time constant (). The filter's time constant is determined by an external timing capacitor (CTIME) attached to the CT pin (pin 4), and an internal current source (IT) connected to CT. The current source is internally fixed at 7.5 A. The resulting time constant in seconds is approximately equal to 3467 times the value of the timing capacitor in Farads. Note that, as a result of the mathematics of RMS detection, the attack and release time constants are fixed in their relationship to each other. The RMS detector is capable of driving large spikes of current into CTIME, particularly when the audio signal input to the RMS detector increases suddenly. This current is drawn from VCC (pin 9), fed through CTIME at pin 4, and returns to the power supply through the ground end of CTIME. If not handled properly through layout and bypassing, these currents can mix with the audio in the circuit's ground structure with unpredictable and undesirable results. As noted in the Applications section, local bypassing from the VCC pin to the ground end of CTIME is strongly recommended in order to keep these currents out of the ground structure of the circuit (see C4 in Figure 2.) The dc output of the detector is scaled with the same constant of proportionality as the VCA gain control: 6.2 mV/dB. The detector's 0 dB reference
(iin0, the input current which causes the detector's output to equal 0V), is trimmed during wafer probe to equal approximately 7.5 A. The RMS detector output stage is capable of sinking or sourcing 125 A. It is also capable of driving up to 150 pF of capacitance. Frequency response of the detector extends across the audio band for a wide range of input signal levels. Note, however, that it does fall off at high frequencies at low signal levels like THAT's other RMS detectors. Differences between the 4305's RMS level
detector circuitry and that of the THAT 2252 RMS detector include the following. 1. The rectifier in the 4305 RMS Detector is internally balanced by design, and cannot be balanced via an external control. The 4305 will typically balance positive and negative halves of the input signal within 10 %, but in extreme cases the mismatch may reach +40 % or -30 % (3 dB). However, even such extreme-seeming mismatches will not significantly increase ripple-induced distortion in dynamics processors over that caused by balanced signal ripple alone. 2. The time constant of the 4305's RMS detector is determined by the combination of an external capacitor CTIME and an internal current source. The internal current source is set to about 7.5 A. A resistor is not normally connected directly to the CT pin on the 4305. 3. The 0 dB reference point, or level match, is also set to approximately 7.5 A. However, as in the 2252, the level match will be affected by any additional currents drawn from the CT pin.
Compressor (or Limiter) Configurations
The 4305 provides the two essential building blocks required for a wide variety of dynamics processing applications. tem. Perhaps the most common application for the 4305 is as a compressor or limiter. These circuits are intended to reduce gain above some determined signal level in order to prevent subsequent stages from being overloaded by too high a signal. Compressors generally have low to moderate compresThe part may be configured into practically any type of dynamics processor sys-
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THAT4305 Pre-trimmed Analog Engine
sion ratios, while limiters have high ratios. In such applications, the signal path has static gain so long as the input signal remains below some threshold, but gain is reduced when the signal rises above the threshold. Compression ratio is defined as the number of dB the input signal increases for a 1 dB increase in output signal.
more details of how the sidechain gain determines compression ratios.
Feedback Topologies
An alternative configuration for compressor/limiter design is to feed the output signal into the RMS detector. The RMS output is fed back (dc coupled) to the VCA's negative control port to reduce signal levels. Similarly as with a feedforward designs, a threshold in the sidechain serves to stop the compression action at low signal levels. The feedback topology behaves somewhat differently from feedforward. First, reaching infinite compression requires infinite gain in the feedback loop from RMS output to VCA control port. Of course, infinite gain is impossible, so practical feedback compressors are usually limited to ratios no greater than 20 or so. Additionally, the gain in the feedback loop alters the effective time constant of the detector, shortening the attack as the ratio becomes higher. This may or may not be appropriate, depending on the desired effect.
Feedforward Topologies
To make a compressor or limiter with a 4305, typically, the input signal is applied to both the VCA and the RMS detector. The RMS output signal is fed forward to the VCA's negative control port (EC-) via a dc-coupled op-amp based stage. This stage has gain above some dc level (the threshold), and no transmission below that level. This path, called the "sidechain," -- from detector output to VCA control port -- determines the compression behavior of the circuit. As signal level rises, the dc voltage at the RMS' output rises. Once the dc level exceeds the threshold, the rms output signal is transmitted through the sidechain and presented to the VCA control port, lowering the gain to signals passing through the VCA. As a result, the output signal level is reduced, or compressed, relative to rising input signal levels. Varying the threshold setting of the sidechain will vary the point at which compression begins. Varying the gain between the RMS output and the VCA control input varies the compression ratio. Feedforward compressor topologies are especially versatile because they cannot become unstable due to oscillation in the control loop. Unity gain in the sidechain produces infinite compression (where the output remains constant regardless of increases in the input signal). With feedforward, negative compression ratios are easily achievable. (Negative compression occurs when the output signal decreases as the input signal increases.) This approximates the effect of playing music backwards, since the attack is suppressed and the release is increased in volume. Many other variations of the feedforward concept are possible. These include implementing more than one threshold, different ratios, additional time constants, ac-coupling of some (or all) of the detector output signal, and many more. See AN101A, The Mathematics of Log-Based Dynamic Processors, for
Expander (Gate) Configurations
By changing the sign of the sidechain in a feedforward compressor, it is possible to arrange signal gain to decrease along with signal level, thus producing an expander. This is typically applied below a threshold (so, the threshold detector's polarity is reversed from that of a compressor) to reduce noise or crosstalk during pauses in program material. This technique has long been used for "cleaning up" individual drum tracks to reduce reverberation, interference from microphones picking up adjacent drum sounds, and alter the attack/decay characteristic of individual drum sounds. Practical gates usually require very fast attack times, and carefully programmable release times. In a 4305, this is best accomplished by using the RMS detector as a log rectifier with very short time constants, and following the detector output with a time-constant stage that applies the desired attack and release behavior. This alters the 4305 detector's natural response characteristics to peak, rather than rms, time constants. We intend to produce an application note showing examples of these circuits. Until that is available, see DN 100, which shows a noise gate application using THAT's 4301 Analog Engine.
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Noise Reduction (Compander) Configurations
An additional application of the 4305 is for noise reduction systems. In these applications, one Analog Engine is configured for use as a compressor to condition audio signals before feeding them into a noisy channel. A second Analog Engine, configured as an expander, is located at the receiver end of the noisy channel. Most commonly, the compression/expansion ratio is modest (e.g. 2:1:2) and is linearly applied across the entire signal dynamic range. During low-level audio passages, the compressor increases signal levels, bringing them up above the noise floor of the channel. At the receiving end, the expander reduces the signal back to its original level, in the process attenuating channel noise. During high-level audio passages, the compressor decreases signal levels, reducing them to fit within the headroom limits of the channel. The expander increases the signal back to its original level. While the channel noise may be increased by this action, in a well-designed compander, at such times the noise floor will be masked by the high-level signal. The 4305 facilitates the design of a wide variety of companding noise reduction systems. The RMS detector responds accurately over a wide range of levels; the VCA responds accurately to a wide range of gain commands; and all the detector and VCA inputs, outputs, and control ports are independently accessible and fully configurable. All these features mean that the 4305 will support a wide range of compander designs, including simple 2:1 wide range (level-independent) systems, level-dependent systems with thresholds and varying compression slopes, systems including noise gating and/or limiting, and systems with varying degrees of pre-emphasis and filtering in both the signal and detector paths. Furthermore, much of this can be accomplished by extensively conditioning the control voltage sidechain rather than the audio signal itself. The audio signal can pass through as little as one VCA and one opamp, and still support multiple ratios, thresholds, and time constants. Note that the 4305 is fully compatible with other Analog Engines from THAT Corporation. All our Analog Engines feature log-responding true-RMS level detectors and exponentially controlled Blackmer
VCAs.
It is possible to compress (encode) signals
using the low-voltage, low-power 4315 or 4320 in a handheld, battery-operated device such as a wireless microphone or instrument belt pack, and expand (decode) that signal using the 4305 in a rack-mount, line-operated receiver.
The Mathematics of Log-Based Dynamics Processors
At first, the logarithmic output of the RMS detector and the exponential control ports of the VCA can be intimidating for designers unfamiliar with THAT Corporation's offerings. However, in fact, these characteristics make developing audio processors easy once a designer understands the concepts involved. As noted earlier, AN101A: The Mathematics of Log-Based Dynamics Processors, discusses these concepts in some detail. The following discussion draws heavily from that application note.
The Feedforward Compressor
Figure 3 shows a conceptual diagram of a very simple feedforward compressor. that Out dB = IndB + GdB , and that GdB = - k IndB . Note that the sign of k makes this a compressor in which gain GdB decreases as input signal level IndB increases. Combining these equations, Out = IndB - k IndB = IndB (1 - k) . Rearranging yields
IndB Out dB
Using the "log
math" principles explained in AN101A, we can state
=
1 (1- k)
= C. R.
This is the compression ratio.
IndB
RMS
IndB
G
-k
dB
Out dB
Figure 3. Simplified feedforward compressor, conceptual diagram.
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THAT4305 Pre-trimmed Low-power Analog Engine
By inspection we can see that if k equals zero, the compression ratio will be 1:1, and if k equals 1, the compression ratio will be infinity:1. Thus, we can make a feedforward compressor/limiter by having the gain of the sidechain vary from zero to one. Note that if k>1, the compression ratio becomes negative. Negative compression results with 1Note that the sign of k makes this an expander in which gain GdB decreases as input signal level IndB decreases. Combining these equations: Out dB = IndB + k IndB , and Out dB = (1 + k)IndB . Rearranging yields
Out dB IndB
The Feedback Compressor
We can use the same approach for a feedback compressor. Figure 4 shows a simplified theoretical model of a feedback compressor. By inspection, Out dB = IndB + GdB , and GdB = - k Out dB . Therefore, Out dB = IndB - k Out dB , and Out dB + k Out dB = IndB . As such,
IndB Out dB
= 1 + k = E. R.
This is the expansion ratio.
IndB
RMS
IndB
G
k
dB
Out dB
= 1 + k = C. R. .
Figure 5. Simplified feedforward expander, conceptual diagram
IndB
G
dB
Out dB
RMS
Out dB
Adjusting the Level Match Point
In the equations so far, we have made the implicit assumption that the decibel reference level everywhere is that of the rms-level detector. This assumption simplifies the math, but it may not correspond to real-world reference levels such as 1V rms (0 dBV) or 0.775V rms (0 dBu). Additionally, it is possible to offset the VCA's inherent behavior of producing unity (0 dB) current gain at 0 mV control voltage (EC+-EC-) by selecting asymmetrical voltage-to-current and current-to-voltage converting resistors (R3 and R2, respectively, in Figure 2). Figure 6 allows for a VCA voltage gain offset of AdB, as well as an offset (LMdB) to vary the "level match" point of the RMS detector. Using similar "log math" from AN101A, we can state for Figure 6: Out dB = IndB + GdB + AdB , where GdB is the VCA's control port gain in dB, and AdB is any static gain or attenuation (in dB) applied to the signal. We can also state that: GdB = - k (Out dB - L. M.dB ) , where L.M. is a varying dc voltage intended to change the system's zero dB reference point (often referred to as level match
-k
Figure 4. Simplified feedback compressor, conceptual diagram.
In this case, as mentioned earlier, infinite compression requires infinite sidechain gain. Fortunately, compression ratios of between 10 and 20 limit effectively enough that infinite gain is not required.
Log-Based Expanders
Similarly, for the feedforward expander shown in Figure 5, we can state that Out dB = IndB + GdB , and that GdB = k IndB .
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Document 600067 Rev 00
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point) from that of the detector, and k is the gain of the sidechain. Substituting the second equation into the first, Out dB = IndB + k(L. M.dB - Out dB ) + AdB , or Out dB = IndB + k x L. M.dB - k x Out dB + AdB , or Out dB =
IndB + k xL.M.dB + AdB (1+ k)
verting resistor (R1 in Figure 2), and b) varying the static signal-path gain offset (AdB). For the 4305, the best practice is to achieve level match via a combination of setting the RMS detector's voltage-to-current conversion, and offsetting the signal-path gain. Signal-path gain can be offset by altering the values of the VCA's voltage-to-current and current-to-voltage resistors. Both of these approaches have the advantage of being temperature-independent. Figure 8 shows the behavior of the compressor of Figure 6 with varying sidechain gain k for a con-
.
IndB
G
dB
A dB
RMS
Out dB
Out dB
stant AdB and LMdB. Note that as k increases, the compression ratio increases, and the dynamic range of output levels decreases. Low input signal levels (including noise) are increased, and high input signal levels are decreased. Figure 9 shows how the same compressor behaves with varying LMdB, but fixed AdB and k. Note that as LMdB is reduced, all output levels decrease, and vice-versa.
-k
+ -
L.M.dB
Figure 6. Feedback compressor with level match
20 LM= -40
Similarly, for the expander shown in Figure 7, we can state that:
dB Out
0 -20 -40 -60 -80 -100 -120 -120 -100 -80 -60 -40 dB In -20 k=0 k=1 k=2 k=3
Out dB = IndB - AdB + GdB GdB = k(IndB - L. M.dB ) . Thus: Out dB = (1 + k)IndB - k x L. M.dB - AdB
0
20
In dB
RMS
IndB
-AdB
+ -
G
k
dB
Out dB
Figure 8. Compressor input-output transfer characteristics with varying k.
0 -20
k=1
L.M.dB
dB Out
-40 -60 -80 -120 -100 -80 -60 -40 dB In L.M.= -10 L.M.= -20 L.M.= -30 L.M.= -40 -20 0 20
Figure 7. Feedforward expander with level match.
In both Figures 6 and 7, we show two ways to adjust the point at which input and output levels of the compressor or expander become equal. These are a) adjusting the 0 dB reference voltage level of the RMS detector by setting its voltage-to current con-
Figure 9. Compressor input-output transfer characteristics with varying LMdB
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THAT4305 Pre-trimmed Low-power Analog Engine
140 100 60 20 -20 -60 -100 -140 -60 -40 dB In -20 0 k=0 k=1 k=2 k=3
40 20 0
dB Out
dB Out
-20 -40 -60 -80 -100 -120 -60 -50 -40 -30 dB In -20 -10 0 L.M.= -10 L.M.= -20 L.M.= -30 L.M.= -40
Figure 10. Expander input-output transfer characteristics with varying k.
Figure 11. Expander input-output transfer characteristics with varying LMdB
Figures 10 and 11 show how the input-output transfer characteristics of the expander of Figure 7 change while varying k (Fig 10) and LM (Fig 11). In each case, all other parameters remain fixed.
So, for signals above this level, GdB = - k(IndB - L. M.- T. A.) + AV . Substituting yields Out dB = IndB - k(IndB - L. M.- T. A.) + AV . For input signals below the level set by the threshold setting, the signal at the output of the ideal diode threshold is 0 (dB), so GdB = AdB , thus, Out dB = IndB + AdB . In the circuit of Figure 12, static gain offset is applied via a dc voltage summed into the sidechain and applied to the VCA control port. This illustrates an alternative method of varying VCA gain (different from offsetting the V-I and I-V converting resistors as mentioned earlier). This is especially convenient when the level match must be varied by the user, as with a front-panel control. Note, however, that since the VCA gain scale factor varies slightly with temperature (+0.33%/C), the level match point will vary
Compressors with Thresholds
The compressor of Figure 6 and expander of Figure 7 form the primary basis for the linear companding systems used in many audio applications. However, they are limited in application, since the compression ratio is linear over the entire dynamic range of the applied signal. While this is fine for companding systems, it is impractical for an effects compressor, not least due to the way very low-level signals -- including noise -- are raised in gain by the compressor's actions. A more practical approach to effects compressors is shown in Figure 12, which offers control over the threshold of compression and gain offset (or "makeup gain") in addition to the ratio. In Figure 12, we've extended this approach to model a compressor with more of the features one would expect in an effects compressor/limiter. This model has a threshold adjustment (TdB), a threshold (set by the ideal diode), a means to vary the sidechain gain (k), and a make-up gain adjustment (AdB). As with the previous equations, all these variables are expressed in decibels, in keeping with the simple "log math" of AN101A. As with all the previous figures, Out dB = IndB + GdB For input signals above the level determined by the threshold setting, IndB > TdB .
IndB
RMS
IndB
GdB
+ Ideal Diode
Out dB
S
-k
S
TdB
A dB
Figure 12. Feedforward compressor with threshold (T), gain (A), and ratio (k) adjustments
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Document 600067 Rev 00
Page 11
slightly with temperature unless the applied voltage is appropriately compensated. If we let TdB=20, AdB=0 dB, and k=0.75, this behavior yields the transfer function shown in Figure 13. As predicted by the above equations, this results in a 4:1 compression ratio above the threshold of -20 dB (relative to the RMS detector's 0 dB reference level. The output level increases by 10 dB over a 40 dB change in input level.
ment (AdB) just as in Figure 8. output. Once again we start with Out dB = IndB + GdB
However, in this
case, the detector level is based on the compressor's
For output signals above the level determined by the threshold setting, Out dB > TdB . So for signals above the threshold, GdB = - k(Out dB - TdB ) + AdB . Substituting yields Out dB = IndB - k(Out dB - TdB ) + AdB ,
20 10 0 -10 -20 -30 -40 -50 -60 -70 -80 -90 -100 -100
dB Out
Out In
which can be reduced to Out dB =
IndB + TdB + AdB (1+ k)
.
For output signals below the level set by the
-80 -60 -40 -20 0 20
threshold setting, the signal at the output of the ideal diode threshold is 0 (dB), so GdB = AdB , and Out dB = IndB + AdB
dB In
Figure 13. Transfer function of a feedforward compressor
In Figure 14, we have again extended the basic feedback compressor with a threshold adjustment (TdB), a threshold (the ideal diode), a means to vary the sidechain gain (k), and a make-up gain adjust-
If we let TdB=10, AdB=20, and k=10, this behavior yields the transfer function shown in Figure 15. The compression ratio of 11:1 allows a rise of only about 4.5 dB over a 50 dB range.
IndB
GdB
RMS Ideal Diode + -
0
Out dB
Out dB
dB In
-50
S
-k
S
Out In
A dB
TdB
-100 -100
-50
0
dB Out
Figure 14. Feedback compressor with threshold, gain, and ratio adjustments
Figure 15. Transfer function of a feedback compressor
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Page 12
THAT4305 Pre-trimmed Low-power Analog Engine
Applications
In this datasheet, we will show detailed circuits for the 4305 in a relatively simple above-threshold compressor, and in two simple companding systems. As mentioned above, many other configurations of the 4305 are possible. THAT intends to publish additional circuits in forthcoming applications notes. Please check with THAT's applications engineering department to see if your application has been covered yet, and for personalized assistance with specific designs. rent at the VCA output, which is converted back to 15 V by U2 and R13 (also 20 k). C2 provides AC coupling, required to block any DC currents that might otherwise flow into the VCA input. This prevents changes in gain from modulating this current, which could produce audible "thumps". The compensation circuit of R28 and C16 is required for stability. The VCA must "see" a source impedance no greater than 5 k above 1 MHz. R28 in parallel with R14 accomplishes this. C16 is chosen to prevent the added noise gain of the lower source impedance from increasing noise within the audio band. Note that such compensation is unnecessary when the voltage-to-current converting resistor (R14) is 5 k or less. For example, if the input signal were limited to lower voltages, the input voltage-to-current converting resistor (R14) could be reduced in value, possibly eliminating the need for R28 and C16. U2, along with C4 and R13 forms a transimpedance amplifier that converts the VCA's output current into a voltage. C4 prevents the VCA's output capacitance from destabilizing the op-amp in this configuration.
Feedforward Compressor/Limiter
The circuit in Figure 16 shows a typical hardknee, feedforward compressor/limiter. In addition to compression ratio, the sidechain includes controls for threshold and make-up gain as well.
The Signal Path
The input of the VCA (pin 15) is a virtual ground, and R14 converts the input signal into a current flowing into the VCA. The maximum total signal current, (IIN + IOUT) is 1.8 mA with 15 V supplies, so R14 is sized to keep the maximum current at unity gain to below this level. With peak input voltage swing limited by the 15 V supply rails, the 20 k resistor at R14 limits maximum iIN to about 750 A. At 0 dB gain, this will cause the same cur+15V C16 100p R28 6k2 R14 20k
15 11 9 EC+ Vcc 13 VCA In ECVee 8 VCA Out Gnd 6
R13 C4 20k
U1A THAT4305
22p NPO
In
C2 10u
Out
U2 C6 22p +15V
-20 dB
-
12
-15V
R12 10k
R8 620k
Gain U1B THAT4305 R1 33k
2 RMS In RMS Out CT 4 5 +20 dB Increase
R18 5k1 C5 100n
C1 10u
R17 5k1
R2 10k D2 D1 1N4148 1N4148 C15 22p
INF:1 CR
-15V R9 1k Compression Ratio
1:1 CR
R3 10k
U4 R11 10k
+15V
C13 22u
CTIME 10u +15V
-40 dBu
R10 10k
R7 430k
Threshold
20 dBu
Increase
U3
-15V
Figure 16. 4305-based feedforward compressor
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Document 600067 Rev 00
Page 13
The Sidechain
As noted earlier, for most effects compressors, it is best not to linearly compress the entire dynamic range of the audio signal. To this end, in the sidechain of figure 16, we have added a threshold amplifier with 30 dB of threshold adjustment. This restricts compression to signals above the threshold, passing those below threshold without any change. Additionally, we added a compression ratio control adjustable from 1:1 to :1. static, or "make-up" gain. The input signal is fed into the RMS detector through C1 and R1. Like the VCA, the input to the detector is AC coupled. This prevents any DC current flowing into the detector's input from being measured by the detector. If unchecked, such offsets would limit resolution at low levels, producing an artificial "floor" to the detector's low-level response. As previously discussed (in the theory section), the output of the detector is proportional to the log of the RMS level of the input voltage. The output of the detector will be approximately zero volts at the "zero dB reference level" -- the point at which the RMS value of the input current equals the timing current (set at 7.5 A for the 4305). We have chosen a value of -10dBu, (245 mVRMS) for the zero dB reference level. The required input resistor can be calculated as R=
245 mVRMS 7 .5A
The scaling at the output of the detector is +6.2 mV/dB, but because R2 is approximately twice R17, the threshold amplifier (U3) has a gain of -2, so the scaling at its output is -12.4 mV/dB. To swing the threshold over 30 dB, we can calculate the required value of R7 as follows:
15 V( 10 k ) R
7 V 0.0124 dB
= 30dB .
Finally,
We can rearrange this to be
10 k R7
we've added a means to apply up to 20 dB of =
30 dB 0.0124 V 15 V dB
, and therefore,
R7 = 403 . 2 k We chose 430 k for R7. U4 is a variable-gain inverter that serves to buffer the VCA's control port, ensuring a lowimpedance drive at that point. distortion at high signal levels.) (High impedances, Above threshold, even as little as 50 to 100 ohms, will increase VCA when U4's gain is -1/2, the net gain of the sidechain (from RMS output to VCA control input) is unity, and the compression ratio is :1. The network of R3, R9, and R11 in conjunction with R18, allows the gain of U4 to vary from 0 to -1/2, and simultaneously shapes the (linear) pot's response so that 50% rotation results in 4:1 compression. 4:1 ratio at 50% rotation is often considered a useful target. Note that with the more conventional approach of connecting the undriven end of the pot to ground, the compres-
= 32. 6 k 33k .
sion ratio at 50% rotation would be 2:1. Finally, R8 and R12 provide the means for add-
Inverting threshold amplifier (U3) provides gain of approximately -2 to the detector output signal above threshold, and zero gain (AV=0) to signals below threshold. The change in gain is accomplished by D1 and D2, which allow negative-going output signals to pass but block positive-going ones. Because U3 is configured to invert, positive-going signals at the RMS output (indicating increasing ac input levels) are passed onwards, while negative-going RMS outputs are blocked. By feeding variable dc into this stage via R7 and the threshold pot R10, we can vary the point at which RMS output signals begin to be passed through to the threshold amplifier stage's output (at the junction of D2 and R2.)
ing static, or "make-up" gain. fore,
1 15 V( 5.R k ) 8 V 0.0062dB
The control-voltage
sensitivity at the output of U4 is 6.2 mV/dB. There-
= 20dB .
We can rearrange this to be
5.1k R8
=
20 dB 0.0062 V 15 V dB 5 k
, and therefore,
R8 =
20 dB V 0.0062dB 15 V
= 625k
We've chosen 620 k for R8 since it is the nearest 5% value.
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Page 14
THAT4305 Pre-trimmed Analog Engine
The signals in the sidechain, and at the output of U4, are generally relatively slow moving, so the sidechain does not usually require wide bandwidth. Furthermore, noise on the VCA control port can modulate the VCA signal, thus adding noise to the signal path. Accordingly, we added C5 in order reduce the noise gain of U4 at high frequencies, which slightly reduces the high-frequency noise floor of the entire circuit. C5 may be omitted for non-critical applications. As described in the Theory of Operation section ("The RMS Detector - In Brief"), the RMS detector is capable of driving large spikes of current into the averaging capacitor CTIME. To prevent these currents from upsetting circuit grounds, it is necessary to bypass VCC to a point very near the grounded end of CTIME with a capacitor equal to or greater than the value of CTIME. This is C13 in Figure 16. The grounded ends of these two capacitors should be connected together before being tied to the rest of the ground system. Doing so will ensure that the current spikes flow within the local loop consisting of the two capacitors, and stay out of the ground system.
a companding system is positioned before the noisy channel (wireless link, storage system, etc.). The static gain of this circuit is 1, or zero dB, and a 5.1 k resistor (R3) along with a 220 pF capacitor (C4) comprise the compensation network is required to keep the VCA's input amplifier stable for all gains. Since the RMS detector output is tied directly to the VCA's EC-, the compression ratio will be 2:1. Note that the use of the negative-sense control port, EC-, makes this circuit a compressor. The RMS detector timing capacitor is set for a release rate of -125 dB per second by using a value of 10 uF. As described in the Theory of Operation section ("The RMS Detector - In Brief"), the RMS detector is capable of driving large spikes of current into the averaging capacitor CTIME. To prevent these currents from upsetting circuit grounds, it is necessary to bypass VCC to a point very near the grounded end of the CTIME with a capacitor (C5) equal to or greater than the value of CTIME. The grounded ends of these two capacitors should be connected together before being tied to the rest of the ground system. Doing so will ensure that the current spikes flow within the local loop consisting of the two capacitors, and stay out of the ground system.
Companding Systems
The Encoder
Figure 17 shows the 4305 configured as a simple 2:1 encoder or feedback compressor. The encoder in
The output of the RMS detector is zero volts when the RMS input current is equal to the timing current (internally set to ~7.5 mA). A voltage level
C5 22p
INPUT
C1 22u
R1 20k R3 5k1 C4 220p
U1A 11 THAT4305 EC+ 13 15 VCA In VCA Out EC12
R2 20k U2A Op-Amp OUTPUT
5 U1B THAT4305
RMS In RMS Out 2 CT 4
+15V
R4 5k1
C2 22u
C3 10u
C5 22u
Figure 17. 4305 simple compander circuit - 2:1 encoder (compressor)
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Document 600067 Rev 00
Page 15
INPUT
C12 22u
+15V
R10 5k1 C13 C14 10u
22u
U3B THAT4305 RMS In 5 2 RMS Out CT 4
C15 U3A THAT4305 11 EC+ 47p R7 20k 13 2 3 U4A 1 Op-Amp OUTPUT
C11 22u
R8 20k R13 5k1 C21 220p
15
VCA In VCA Out EC12
Figure 18. 4305 Typical Application Circuit - 1:2 Expander
column labeled RMS In are derived using the equaof -28.5 dBV was chosen as the desired zero dB reference. The RMS detector's input resistance can be calculated as: R RMS in =
-28. 5 10 20
tion: I RMS In =
10
( Encoder Out ) 20
RRMS In
7 .5A
5 .1 k .
The required encoder VCA gain range is -24 dB to +36 dB, and the required decoder VCA gain range is -36 dB to +14 dB. These gains are easily within the capabilities of the 4305's VCA. The range of RMS input currents is easily accommodated at the high end, though accuracy may be slightly compromised at the lowest input levels.
Encoder In (dBV) +20 +10 0 -10 -20 -30 -40 -50 -60 Encode VCA Gain (In dB) -24 -19 -14 -9 -4 1 6 11 16 21 26 31 36 Encoder Out/ Decoder In (dBV) -4 -9 -14 -19 -24 -29 -34 -39 -44 -49 -54 -59 -64 IRMS In (mA) 0.1223 0.0688 0.0387 0.0218 0.0122 0.0069 0.0039 0.0022 0.0012 0.0007 0.0004 0.0002 0.0001 Decoder VCA Gain (In dB) 24 19 14 9 4 -1 -6 -11 -16 -21 -26 -31 84 Decoder Out (dBV) 20 10 0 -10 -20 -30 -40 -50 -60 -70 -80 -90 139
This value also applies to the decoder.
The Decoder
Figure 18 shows the THAT4305 configured as a 1:2 expander intended to complement the encoder in Figure 17. This circuit also uses a static gain of zero dB. Since the VCA is not stable unless it sees a high frequency source impedance of 5 k or less, the compensation network of R13 and C21 ensures stability. In this instance, the RMS detector output is connected to EC+; this reverses the polarity of the control signal relative to the encoder, and makes this circuit a 2:1 expander.
System Performance
Table 3 shows the transfer characteristics of this companding system. The columns labeled Encoder VCA Gain, Encoder Out, Decoder VCA Gain, and Decoder Out use the equations derived previously in the Theory sub-section entitled "The Mathematics of Log Based Companding Systems". The values in the
-70 -80 -90 -100
Table 3. 2:1 compander transfer characteristics
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Page 16
THAT4305 Pre-trimmed Analog Engine
Figure 19 tracks signal levels through the encoder and decoder of Figures 17 and 18. The encoder reduces the dynamic range at its input by a factor of 2, compressing 120dB into 60dB. The decoder expands this dynamic range back to track that of the encoder's input signal.
Compression Process Expansion Process
given signal level, high-frequency signals are lowered in level by the VCA more than low-frequency signals. As an additional enhancement, we have included a means to truncate the RMS detector's low-level response. This improves low level tracking between different detectors by forcing each detector to "bottom out" at a predetermined level, eliminating the effects of different low-level behavior from one detector to the next.
20
System Performance
0 -20 20 0 -20 -40 -60 -80 -100
The compander shown in Figures 20 and 21 implements all of the aforementioned improvements. Assuming no change in VCA gain (GdB), the pre-emphasis network of R3 and C7 produces ~20 dB of signal-path pre-emphasis starting at ~2 kHz and stopping at ~19 kHz. Note that R3 and C7 also compensate the input to the VCA, so additional components are not required to implement this feature. Signal fed to this network is buffered by U2; while this buffer is not always necessary, the pre-emphasis network must be driven from a low source impedance to ensure proper tracking between the encoder pre-emphasis and the decoder de-emphasis. If driven from an unbuffered source, the pre-emphasis network should be adjusted to take into account the impedance of that source. We have included ~10 dB of RMS pre-emphasis (provided by R5 and C8 in the encoder, and R11 and C18 in the decoder) for the detectors in both the encoder and the decoder. The center frequency of this pre-emphasis circuit is aligned with the center frequency of the signal path pre-emphasis when evaluated on a logarithmic frequency scale. This shifts the level match of the encoder symmetrically about the mid-point of the signal-path pre-emphasis, which configures the system to take the best advantage of the companding to avoid high-level high-frequency overload in the transmission or storage channel. R6 of the Hi-Fi encoder and R12 of the decoder
dB
-40 -60 -80
-100 In(Cmp) Out Cmp In(Exp) Out Exp
Figure 19. 2:1 compander transfer characteristics
Hi-fi Compander
While the previous circuits perform adequately in some applications, a few minor changes can result in substantially improved overall performance. The following compander implementation adds preand de-emphasis to the signal path. Signal path pre-emphasis helps overcome the rising noise level with frequency of an FM RF channel by raising the level of the high frequency portions of the signal before it passes through the transmission channel. Matching signal-path de-emphasis in the decoder brings the frequency response back to flat while simultaneously lowering the noise floor of the channel. This helps ensure that isolated low-frequency signals mask the channel noise by reducing the perception of high-frequency noise signals. Of course, the drawback of signal-path
are intended to force each of the detectors to stop responding to low level signals at the same point in order to improve tracking. This floor occurs when the RMS current through R1 equals that of R3, and when the current through R10 equals that of R12. Since the input of the RMS detector is at virtual ground, the current through R3 and R12 will be
pre-emphasis is that it can cause overload in the channel when high-level, high-frequency signals are present. To guard against this problem, we have added RMS pre-emphasis to both detectors. level-match point to high-frequency signals. This For a mitigates high-frequency overload by lowering the
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Document 600067 Rev 00
Page 17
i R3 = i R12 =
RMSOut R3 RMSOut R12
, and
VIn = 10
( -52dBV ) 20
= 316mVRMS . Therefore, . , And
0.00316 mV 8.87 k
=
( -23.6 -( -52)) 0.0062 R3
We'll choose a point about 4dB above the minimum input level which is -56dBV.
R3 =
( -23.6 -( -52)) 0.0062
0. 00316 mV 8. 87 k
= 459k 464 k
The same is true for R12.
Signal Path Pre-emphasis C7 Encode In U2 3n3 C1 1u R3 2k32 R1 21k0 EC12 R6 464k RMS Out RMS In 2 CT 4 C3 10u R5 15 11 EC+ U1A THAT4305 13
C5 22p R2 84k5 U3 Op-Amp C8 Encode Out
VCA In VCA Out
Op-Amp
4k02 3n3 RMS Pre-emphasis R4 8k87 C2 1u
5 Vcc C4 22u U1B THAT4305
Figure 20. 4305 hi-fi 2:1 encoder circuit
RMS Pre-emphasis
C18 3n3 Decode In U4 C12 1u Op-Amp V+ C14 22u C11 1u R8 84k5
R11 4k02
R12 464k U3B THAT4305 RMS In 5 2 RMS Out CT 4
Signal Path De-emphasis
R10 8k87
R9 2k32 U3A THAT4305 11 EC+ R7 21k0 U5
C17 3n3 C15 22p Decode Out
C13 10u 15 R13 5k1 C21 220p
VCA In VCA Out EC12
13
Op-Amp
Figure 21. 4305 hi-fi 2:1 decoder circuit
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Page 18
THAT4305 Pre-trimmed Analog Engine
Encoder In (dBV) 20 10 0 -10 -20 -30 -40 -50 -60 -70 -80 -90 -100
Encoder VCA Gain (In dB) -28 -23 -18 -13 -8 -3 2 7 12 17 22 27 32
Encoder Out/ Decoder In (dBV) 4 -1 -6 -11 -16 -21 -26 -31 -36 -41 -46 -51 -56
IRMS In (mA) 0.1841 0.1035 0.0582 0.0327 0.0184 0.0104 0.0058 0.0033 0.0018 0.0010 0.0006 0.0003 0.0002
Decoder VCA Gain (In dB) 28 23 18 13 8 3 -2 -7 -12 -17 -22 -27 -32
Decoder Out (dBV) 20 10 0 -10 -20 -30 -40 -50 -60 -70 -80 -90 -100
ratios other (higher or lower) than 2:1:2, multi-band companders, etc. The 4305 is versatile enough to be used as the heart of a compressor, expander, noise gate, AGC, de-esser, frequency-sensitive compressor, and many other dynamics processors. It is beyond the scope of this data sheet to provide specific advice about any of these functional classes. We refer the interested reader to THAT's applications notebooks volumes 1 and 2, which contain many circuits based on THAT's other VCAs and RMS level detectors, but are largely applicable to the 4305 with only minor variations. Of course, look for more applications information aimed specifically at the 4305 in the future.
Compression Process Expansion Process
Table 4. Hi-fi compander transfer characteristics
20 0 -20
Table 4 shows the transfer characteristics of this companding system (neglecting the effects of R6 and R12). As before, the columns labeled Encoder VCA Gain, Encoder Out, Decoder VCA Gain, and Decoder
dB
Out use the equations derived previously in the section titled "The Mathematics of Log Based Companding Systems". The values in the column labeled RMS In are derived using the equation: I RMS In =
10
( Encoder Out ) 20
-40 -60 -80
20 0 -20 -40 -60 -80 -100
RRMS In
-100 In(Cmp) Out Cmp In(Exp) Out Exp
Figure 22 tracks signal levels through the encoder and decoder of Figures 20 and 21. The compression and expansion ratios here are the same as those of the previous circuits, but the frequency shaping afforded by signal pre- and de-emphasis and detector pre-emphasis make this a superior sounding system. In this application, the VCA gain ranges over about 30 dB, which is well within specification, as is the RMS detector input current.
Figure 22. Hi-fi compander transfer characteristics
Closing Thoughts
The design of dynamics processors and companding systems is a very intricate art: witness the proliferation of dynamics processors available in the market today. Many of these are based on THAT's VCAs and level detectors, yet they all have individual sonic characteristics. In the applications section of this data sheet, we have offered a few examples only as starting points. THAT Corporation's applications engineering department is ready to assist customers with suggestions for tailoring and extending these basic circuits to meet specific needs.
Other Dynamics Processor Configurations
We have said before that the building blocks contained within the 4305 are applicable to a very wide range of dynamics processor configurations. These include companding noise reduction systems with
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Document 600067 Rev 00
Page 19
Package and Soldering Information
The THAT 4305 is available in a 16-pin QSOP package. The package dimensions are shown in Figure 23 below, while the pinout is given in Table 1 on page 1. The 4305 is available only in a lead-free, "green" package. The lead frame is copper, plated with successive layers of nickel palladium, and gold. This approach makes it possible to solder these devices using lead-free and lead-bearing solders. The plastic mold compound, and the material in which the parts are packaged, contains no hazardous substances as specified in the RoHS directive. For more information, including MDDS forms which disclose the substances contained in our ICs and their packaging, please visit www.thatcorp.com/RoHShome.html. The package has been qualified using reflow temperatures as high as 260C for 10 seconds. This makes them suitable for use in a 100% tin solder process. Furthermore, the 4305 has been qualified to a JEDEC moisture sensitivity level of MSL1. No special humidity precautions are required prior to flow soldering the parts.
1 D A
E B C J H G
I
ITEM A B C D E G H I J
0-8
INCHES 0.189 - 0.196 0.150 - 0.157 0.228 - 0.244 0.008 - 0.012 0.025 BSC 0.0532 - 0.0688 0.004 - 0.010 0.016 - 0.050 0.0075 - 0.0098
MILLIMETERS 4.80 - 4.98 3.81 - 3.99 5.79 - 6.20 0.20 - 0.30 0.635 BSC 1.35 - 1.75 0.10 - 0.25 0.40 - 1.27 0.19 - 0.25
Figure 23. QSOP-16 surface mount package drawing
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Page 20
THAT4305 Pre-trimmed Analog Engine
Notes:
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